Class AB output stage

ABSTRACT

This disclosure describes at least one class AB amplifier output stage circuit arrangement that can operate at low supply voltages, with minimum current generated. Furthermore, at least one class AB amplifier stage circuit arrangement described herein reacts favorably to a supply voltage, that is, exhibits a good power supply rejection ratio. Moreover, this disclosure describes class AB amplifier output stage circuit arrangements that include a negative channel metal oxide semiconductor (NMOS) transistor current mirror arrangement and a positive channel metal oxide semiconductor (PMOS) transistor current mirror arrangement. In some implementations, a monitoring circuit may be coupled to a class AB amplifier output stage circuit arrangement to offset mismatch that may occur in the class AB amplifier output stage.

RELATED APPLICATIONS

This Application is a Divisional Application of U.S. application Ser.No. 12/367,989, which was filed on Feb. 9, 2009. The entire contents ofU.S. application Ser. No. 12/367,989 are hereby incorporated herein byreference.

BACKGROUND

Several different amplifier circuit arrangements may be utilized toprovide output signals. In one example, a class A amplifier circuitarrangement reproduces an entire input signal because an active elementof the class A amplifier circuit arrangement, such as a transistor, isconstantly in the active mode. However, class A amplifiers typicallyhave a high power consumption because the active element does not stopconducting current.

In another example, a class B amplifier circuit arrangement reproduceshalf of the input signal since an active element of a class B amplifiercircuit arrangement spends half of the time in active mode and the otherhalf in cutoff. Class B amplifier circuit arrangements may include apush-pull configuration that has two active elements with one activeelement in active mode for half of an input waveform and the otheractive element in active mode for the other half of the input waveform.The properties of class B amplifier circuit arrangements may vary withload conditions and may suffer from harmonic distortion when the handofffrom one element to another does not occur properly.

Class AB amplifier circuit arrangements are a mixture between class Aamplifier circuit arrangements and class B amplifier circuitarrangements. Class AB amplifier circuit arrangements include two activeelements that are in the active mode more than 50% of the time todecrease the amount of harmonic distortion that occurs during thehandoff from one active element to another. Adjustment of outputquiescent current in Class AB amplifier circuit arrangements may beproblematic.

Some class AB amplifier circuit arrangements may utilize a stack of twodiodes to adjust quiescent current. However, such arrangements do nottypically operate well at low supply voltages because the stack of twodiodes requires a relatively high supply voltage. Other class ABamplifier circuit arrangements may operate at low supply voltages, butmay have increased current consumption. For example, class AB amplifiercircuit arrangements may include current mirrors that generate a currentinternally and mirror the current to the output of the circuitarrangement. However, the current consumption of such class AB amplifiercircuit arrangements is doubled. Still other class AB amplifier circuitarrangements may operate at low supply voltages and have low currentconsumption, but suffer from a poor power supply rejection ratio.

BRIEF DESCRIPTION OF THE DRAWINGS

The detailed description is described with reference to the accompanyingfigures. In the figures, the left-most digit(s) of a reference numberidentifies the figure in which the reference number first appears. Theuse of the same reference number in different instances in thedescription and the figures may indicate similar or identical items.

FIG. 1 is a schematic diagram of an amplifier circuit arrangementutilized to provide an amplified input signal from a source to a load.

FIG. 2 is a schematic diagram of a class AB amplifier output stagecircuit arrangement including a PMOS transistor current mirrorarrangement and an NMOS transistor current mirror arrangement.

FIG. 3 is a schematic diagram of a monitoring circuit of a class ABamplifier output stage, where the monitoring circuit includes anoperational transconductance amplifier arrangement.

FIG. 4 is a schematic diagram of a monitoring circuit of a class ABamplifier output stage, where the monitoring circuit includes adifferential amplifier arrangement.

FIG. 5 is a schematic diagram of a monitoring circuit of a class ABamplifier output stage, where the monitoring circuit includes adifferential difference amplifier.

FIG. 6 is a flow diagram of a method of correcting mismatch in a classAB amplifier output stage.

DETAILED DESCRIPTION

This disclosure describes at least one class AB amplifier output stagecircuit arrangement that can operate at low supply voltages, withminimum current generated. Furthermore, at least one class AB amplifierstage circuit arrangement described herein reacts favorably to a supplyvoltage, that is, exhibits a good power supply rejection ratio.Moreover, this disclosure describes class AB amplifier output stagecircuit arrangements that include a negative channel metal oxidesemiconductor (NMOS) transistor current mirror arrangement and apositive channel metal oxide semiconductor (PMOS) transistor currentmirror arrangement. In some implementations, a monitoring circuit may becoupled to a class AB amplifier output stage circuit arrangement tooffset mismatch that may occur in the class AB amplifier output stage.

According to one exemplary implementation, an apparatus includes a firstcurrent mirror arrangement coupled to a first input signal arrangement.The first input signal arrangement includes a first input current sourceand a first impedance. The apparatus also includes a second currentmirror arrangement coupled to a second input signal arrangement. Thesecond input signal arrangement includes a second input current sourceand a second impedance. The first current mirror arrangement is coupledto the second current mirror arrangement.

According to another implementation, an apparatus includes a firstoperational transconductance amplifier including an output node coupledto a first variable current source. The first operationaltransconductance amplifier determines a voltage drop of a firstimpedance. The apparatus also includes a second operationaltransconductance amplifier including an output node coupled to a secondvariable current source. The second operational transconductanceamplifier determines a voltage drop of a second impedance. The outputnode of the second operational transconductance amplifier is coupled tothe output node of the first operational transconductance amplifier.

According to another implementation, an apparatus includes adifferential amplifier arrangement coupled to a first impedance of aclass AB amplifier output stage and coupled to a second impedance of theclass AB amplifier output stage. An output node of the differentialamplifier arrangement is coupled to a variable current sourcearrangement and one or more input nodes of the differential amplifierarrangement are coupled to the first impedance and the second impedance.

According to another implementation, a method includes generating anoutput signal of a first amplifier circuit based on a voltage drop of afirst impedance of a class AB amplifier output stage. The method alsoincludes generating an output signal of a second amplifier circuit basedon a voltage drop of a second impedance of a class AB amplifier outputstage. The method further includes generating a compensating currentwhen the voltage drop of the first impedance is different from thevoltage drop of the second impedance. The compensating current adjuststhe voltage drop of the first impedance, the voltage drop of the secondimpedance, or a combination thereof, such that the voltage drop of thefirst impedance and the voltage drop of the second impedance areadjusted to be approximately equal.

FIG. 1 is a schematic diagram of an apparatus 100 utilized to provide anamplified input signal from a source 102 to a load 106 via an amplifierdevice 104. In particular implementations, the source 102 may includeone or more circuit arrangements that provide one or more input signalsto the amplifier device 104. The input signals may include radiofrequency signals, audio signals, digital signals, or other signalscarrying data. The load 106 may include an additional device thatreceives the output of the amplifier device 104 as an input signal. Forexample, the load 106 may include an output device, such as an audiospeaker, an analog to digital conversion circuit, a mixer, or acombination thereof. In some implementations, the circuit 100 may beincluded in a high speed amplifier.

The amplifier device 104 includes an input stage 108 and a class ABamplifier output stage 110. In an illustrative implementation, the inputstage 108 may include one or more devices, such as operationalamplifiers, to modify signals from the source 102 and provide themodified signals to the class AB amplifier output stage 110. In turn,the class AB amplifier output stage 110 may further modify the signalsfrom the source 102 and provide these signals to the load 106. In aparticular implementation, the class AB amplifier output stage 110 mayinclude a positive channel metal oxide semiconductor (PMOS) transistorcurrent mirror arrangement and a negative channel metal oxidesemiconductor (NMOS) transistor current mirror arrangement. Animpedance, such as a resistor or an arrangement of transistors, may becoupled between the transistors of the PMOS current mirror arrangement.An additional impedance may be coupled between the transistors of theNMOS current mirror arrangement. The circuit arrangement of the class ABamplifier output stage 110 is configured to provide good regulation ofoutput quiescent current, while operating at low supply voltages or highsupply voltages, with a good power supply rejection ratio, and minimumcurrent consumption.

In some implementations, a monitoring circuit 112 may be utilized in theamplifier device 104 to reduce mismatch in the class AB amplifier outputstage 110. In an illustrative implementation, the monitoring circuit 112may include a first operational transconductance amplifier (OTA) togenerate an output current based on a voltage drop of a first impedancecoupled between transistors of a PMOS current mirror arrangement of theclass AB amplifier output stage 110. The monitoring circuit 112 may alsoinclude a second OTA to generate an output current based on a voltagedrop of a second impedance coupled between transistors of an NMOScurrent mirror arrangement of the class AB amplifier output stage 110.The output currents of the first OTA and the second OTA of themonitoring circuit 112 may be utilized to produce a compensating currentto regulate the voltage drops of the impedances of the current mirrorarrangements of the class AB amplifier output stage 110, such that thevoltage drops are approximately equal, in order to provide good controlof output quiescent current of the class AB amplifier output stage 110.

In some instances, the impedance values of the first impedance and thesecond impedance are approximately equal and the voltage drop of thefirst impedance and the voltage drop of the second impedance may beapproximately equal when the first impedance and the second impedancereceive approximately the same current. In other instances, theimpedance values of the first impedance and the second impedance may bedifferent and the current provided to the first impedance and the secondimpedance may correspond to the difference between the impedance values,such that the voltage drop of the first impedance and the voltage dropof the second impedance are approximately equal. For example, when theimpedance value of the first impedance is larger than that of the secondimpedance, the current provided to the first impedance would be lessthan the current provided to the second impedance.

In another illustrative implementation, the monitoring circuit 112 mayinclude a differential amplifier arrangement to offset mismatch in theclass AB amplifier output stage 110. For example, the monitoring circuit112 may include a first differential amplifier to determine a voltagedrop of a first impedance coupled between transistors of a PMOS currentmirror arrangement of the class AB amplifier output stage 110. Themonitoring circuit 112 may also include a second differential amplifierto determine a voltage drop of a second impedance coupled betweentransistors of an NMOS current mirror arrangement of the class ABamplifier output stage 110. The first differential amplifier and thesecond differential amplifier may be coupled to a third differentialamplifier that drives one or more auxiliary variable current sources toproduce a compensating current when there is a difference between thevoltage drop of the first impedance and the voltage drop of the secondimpedance. In an alternative implementation, the differential amplifierarrangement may include a differential difference amplifier thatproduces an output that drives the one or more auxiliary variablecurrent sources to produce a compensating current when there is adifference between the voltage drop of the first impedance and thevoltage drop of the second impedance.

FIG. 2 is a schematic diagram of a class AB amplifier output stagecircuit arrangement 200 including a PMOS transistor current mirrorarrangement and an NMOS transistor current mirror arrangement. The PMOStransistor current mirror arrangement includes a first positive channelmetal oxide semiconductor (PMOS) transistor 202 and a second PMOStransistor 204. The PMOS transistor current mirror arrangement iscoupled to a first input signal arrangement 206. The first input signalarrangement 206 includes a first impedance 208 and a first input currentsource 210. A gate of the first PMOS transistor 202 is coupled to thefirst impedance 208 and a drain of the first PMOS transistor 202 iscoupled to a reference current source 212. A source of the first PMOStransistor 202 is coupled to a supply voltage, V_(DD). A gate of thesecond PMOS transistor 204 is coupled to the first impedance 208 and tothe first input current source 210 and a drain of the second PMOStransistor 204 is coupled to an output terminal 214. Additionally, asource of the second PMOS transistor 204 is coupled to the supplyvoltage V_(DD).

The NMOS transistor current mirror arrangement includes a first negativechannel metal oxide semiconductor (NMOS) transistor 216 and a secondNMOS transistor 218. The NMOS transistor current mirror arrangement iscoupled to a second input signal arrangement 220. The second inputsignal arrangement 220 includes a second impedance 222 and a secondinput current source 224. A gate of the first NMOS transistor 216 iscoupled to the second impedance 222 and a drain of the first NMOStransistor 216 is coupled to the reference current source 212. A sourceof the first NMOS transistor 216 is coupled to a ground 226. A gate ofthe second NMOS transistor 218 is coupled to the second impedance 222and to the second input current source 224. A drain of the second NMOStransistor 218 is coupled to the output terminal 214 and a source of thesecond NMOS transistor 218 is coupled to the ground 226.

In some implementations, the reference current source 212 may bereplaced by two separate matched current sources to provide appropriatebias currents to the PMOS transistor current mirror arrangement and theNMOS transistor current mirror arrangement. For example, one of thematched current sources may drive the first PMOS transistor 202 and theother matched current source may drive the first NMOS transistor 216.

In an illustrative implementation, the first input current source 210,the second input current source 224, or a combination thereof, mayproduce a negative input current decreasing the voltage at the gate ofthe second PMOS transistor 204 and decreasing the voltage at the gate ofthe second NMOS transistor 218. In response to a negative input current,the second PMOS transistor 204 may deliver an increased output currentby producing an increased source to drain current that provides acurrent path between the supply voltage V_(DD) and the output terminal214. The output current of the second NMOS transistor 218 decreases inresponse to a negative input current. In some instances, such as with alarge input current, the output current of the second NMOS transistor218 is shut off and the second NMOS transistor 218 does not conduct anycurrent. In this way, the output transistors 204 and 218 exhibit thedesired push-pull behavior based on a negative input current.

In another illustrative implementation, the first input current source210, the second input current source 224, or a combination thereof, mayproduce a positive input current increasing the voltage of the gate ofthe second PMOS transistor 204 and increasing the voltage of the gate ofthe second NMOS transistor 218. In response to a positive input current,the second NMOS transistor 218 delivers an increased output current,while the output current of the second PMOS transistor 204 decreases. Insome instances, such as with a large input current, the output currentof the second PMOS transistor 204 is shut off and does not conductcurrent from the supply voltage V_(DD) to the output terminal 214. Inthis way, the output transistors 204 and 218 exhibit the desiredpush-pull behavior based on a positive input current.

In a further illustrative implementation, a quiescent condition of theAB amplifier output stage circuit arrangement 200 occurs when no currentis provided by the first input current source 210 or the second inputcurrent source 224. In the quiescent condition, the current provided bythe reference current source 212 is mirrored by the second PMOStransistor 204 and the second NMOS transistor 218. Under somecircumstances, the first input current source 210, the second inputcurrent source 224, or a combination thereof, may produce some currentduring the quiescent condition, which may cause a mismatch condition.

FIG. 3 is a schematic diagram of a monitoring circuit 300 of a class ABamplifier output stage circuit arrangement, such as the class ABamplifier output stage circuit arrangement 200 of FIG. 2. The monitoringcircuit 300 may be configured to offset mismatch that occurs in theclass AB amplifier output stage circuit arrangement. A class ABamplifier output stage circuit arrangement including larger transistorsand receiving an accurate steering current may have minimal mismatch;however, when transistors of a class AB amplifier output stage circuitarrangement are smaller in size, such as transistors of a high-speedamplifier, mismatch may have a detrimental effect on control of thequiescent current.

The monitoring circuit 300 includes a first impedance 302 of a class ABamplifier output stage. In some implementations, the first impedance 302may correspond to the impedance 208 of FIG. 2. A first node of the firstimpedance 302 may be coupled to a low-pass filter 304 and a second nodeof the first impedance 302 may be coupled to a low-pass filter 306. Inaddition, the first node and the second node of the first impedance 302are coupled to a first auxiliary variable current source 308. In someimplementations, the first auxiliary variable current source 308 may beconnected in parallel with the first current source 210 of FIG. 2. Inother implementations, the first auxiliary variable current source 308and the first current source 210 of FIG. 2 may represent the samecurrent source. For example, an output current of the monitoring circuit300 and an input current to the class AB amplifier output stage 200 maybe combined by a summer circuit (not shown), such as an op-amp, and fedinto a single current source 308/210 that provides current across thefirst impedance 302/208.

The monitoring circuit 300 also includes a second impedance 310 of aclass AB amplifier output stage. In some implementations, the secondimpedance 310 may correspond to the impedance 222 of FIG. 2. A firstnode of the second impedance 310 may be coupled to a low-pass filter 312and a second node of the second impedance 310 may be coupled to alow-pass filter 314. The first node and the second node of the secondimpedance 310 are also coupled to a second auxiliary variable currentsource 316. In some implementations, the second auxiliary variablecurrent source 316 may be connected in parallel with the second currentsource 224 of FIG. 2. In other implementations, the second auxiliaryvariable current source 316 and the second current source 224 of FIG. 2may represent the same current source. For example, an output current ofthe monitoring circuit 300 and an input current to the class ABamplifier output stage 200 may be combined in a summer circuit (notshown), such as an op-amp, and fed into a single current source 316/224that provides current across the second impedance 310/222. Further, inparticular implementations, such as with linear circuits and small inputsignals, the low-pass filters 304, 306, 312, 314 may not be necessary.

Additionally, the monitoring circuit 300 includes a first operationaltransconductance amplifier (OTA) 318. A non-inverting node of the firstOTA 318 is coupled to the low-pass filter 306 and an inverting node ofthe first OTA 318 is coupled to the low-pass filter 304. An output nodeof the first OTA 318 is coupled to the first auxiliary variable currentsource 308.

Further, the monitoring circuit 300 includes a second operationaltransconductance amplifier (OTA) 320. A non-inverting node of the secondOTA 320 is coupled to the low-pass filter 312 and an inverting node ofthe second OTA 320 is coupled to the low-pass filter 314. An output nodeof the second OTA 320 is coupled to the second auxiliary variablecurrent source 316. In addition, the output node of the second OTA 320is coupled to the output node of the first OTA 318.

In an illustrative implementation, the first OTA 318 may determine avoltage drop of the first impedance 302 and the second OTA 320 maydetermine a voltage drop of the second impedance 310. The output currentof the first OTA 318 is related to the voltage drop of the firstimpedance 302 and the output current of the second OTA 320 is related tothe voltage drop of the second impedance 310. In some implementations,the voltage drop of the first impedance 302 and the voltage drop of thesecond impedance 310 are approximately equal. When the voltage drop ofthe first impedance 302 and the voltage drop of the second impedance 310are approximately equal, the respective output currents of the first OTA318 and the second OTA 320 cancel each other and have a sum of zero orapproximately zero.

In other implementations, the voltage drop of the first impedance 302may be different from the voltage drop of the second impedance 310. Forexample, the voltage drop of the first impedance 302 and the voltagedrop of the second impedance 310 may be different due to differingcurrents across the first impedance 302 and the second impedance 310 ofthe class AB amplifier output stage. To illustrate, during a quiescentcondition of the class AB amplifier output stage, a first current sourceof the class AB amplifier output stage may produce a slightly positivecurrent and/or a second current source of the class AB amplifier outputstage may produce a slightly negative current. Consequently, animpedance coupled to the first current source would have a differentvoltage drop than an impedance coupled to the second current sourceresulting in mismatch of the class AB amplifier output stage.

When the voltage drop of the first impedance 302 and the voltage drop ofthe second impedance 310 are different, the output current of the firstOTA 318 and the output current of the second OTA 320 drive the firstauxiliary variable current source 308, the second auxiliary variablecurrent source 316, or a combination thereof, to generate a compensatingcurrent, such that the voltage drop of the first impedance 302 and thevoltage drop of the second impedance 310 are adjusted to beapproximately equal. Thus, the monitoring circuit 300 relies on a closedregulation loop to equalize the voltage drop of the first impedance 302and the voltage drop of the second impedance 310, so that the class ABamplifier output stage will avoid the effect of mismatch andsubsequently produce a correct output current as designed.

In some implementations when the voltage drop of the first impedance 302and the voltage drop of the second impedance 310 are different, thefirst auxiliary variable current source 308 and the second auxiliaryvariable current source 316 may receive the combined output currents ofthe first OTA 318 and the second OTA 320 and produce respectivecompensating currents such that the voltage drop of the first impedance302 and the voltage drop of the second impedance 310 are adjusted to beapproximately equal. For example, when a slightly negative current isprovided to the first impedance 302 during the quiescent condition, thefirst auxiliary variable current source 308 may provide a slightlypositive compensating current. Moreover, when a slightly positivecurrent is provided to the first impedance 302 during the quiescentcondition, the first auxiliary variable current source 308 may provide aslightly negative compensating current. In another example, when aslightly positive current is provided to the second impedance 310 duringthe quiescent condition, the second auxiliary variable current source316 may provide a slightly negative compensating current. Furthermore,when a slightly negative current is provided to the second impedance 310during the quiescent condition, the second auxiliary variable currentsource 316 may provide a slightly positive compensating current.

In other implementations, the monitoring circuit 300 may include oneauxiliary variable current source rather than the first auxiliaryvariable current source 308 and the second auxiliary variable currentsource 316 as shown in FIG. 3. For example, the monitoring circuit 300may include an auxiliary variable current source to provide compensatingcurrent to the first impedance 302. In this example, when the voltagedrop of the first impedance 302 and the voltage drop of the secondimpedance 310 are different, the auxiliary variable current sourceprovides a compensating current to the first impedance 302 based on theoutput signals of the first OTA 318 and the second OTA 320, such thatthe voltage drop of the first impedance 302 and the voltage drop of thesecond impedance 310 are adjusted to be approximately equal. In anotherexample, the monitoring circuit 300 may include an auxiliary variablecurrent source to provide compensating current to the second impedance310. In this example, when the voltage drop of the first impedance 302and the voltage drop of the second impedance 310 are different, theauxiliary variable current source provides a compensating current to thesecond impedance 310 based on the output signals of the first OTA 318and the second OTA 320, such that the voltage drop of the firstimpedance 302 and the voltage drop of the second impedance 310 areadjusted to be approximately equal.

FIG. 4 is a schematic diagram of a monitoring circuit 400 of a class ABamplifier output stage circuit arrangement, such as the class ABamplifier output stage circuit arrangement 200 of FIG. 2. The monitoringcircuit 400 may be configured to offset mismatch that occurs in theclass AB amplifier output stage circuit arrangement.

The monitoring circuit 400 includes a first impedance 402 of a class ABamplifier output stage. In some implementations, the first impedance 402may correspond to the first impedance 208 of FIG. 2. A first node of thefirst impedance 402 may be coupled to a low-pass filter 404 and a secondnode of the first impedance 402 may be coupled to a low-pass filter 406.In addition, the first node and the second node of the first impedance402 are coupled to a first auxiliary variable current source 408. Insome implementations, the first auxiliary variable current source 408may be connected in parallel with the first current source 210 of FIG.2, while in other implementations the first auxiliary variable currentsource 408 and the first current source 210 of FIG. 2 may represent thesame current source.

The monitoring circuit 400 also includes a second impedance 410 of aclass AB amplifier output stage. In some implementations, the secondimpedance 410 may correspond to the second impedance 222 of FIG. 2. Afirst node of the second impedance 410 may be coupled to a low-passfilter 412 and a second node of the second impedance 410 may be coupledto a low-pass filter 414. The first node and the second node of thesecond impedance 410 are also coupled to a second auxiliary variablecurrent source 416. In some implementations, the second auxiliaryvariable current source 416 may be connected in parallel with the secondcurrent source 224 of FIG. 2, while in other implementations the secondauxiliary variable current source 416 and the second current source 224of FIG. 2 may represent the same current source. Further, in particularimplementations, the low-pass filters 404, 406, 412, 414 may not benecessary.

Additionally, the monitoring circuit 400 includes a first differentialamplifier 418. A non-inverting node of the first differential amplifier418 is coupled to the low-pass filter 406 and an inverting node of thefirst differential amplifier 418 is coupled to the low-pass filter 404.Further, the monitoring circuit 400 includes a second differentialamplifier 420. An inverting node of the second differential amplifier420 is coupled to the low-pass filter 412 and a non-inverting node ofthe second differential amplifier 420 is coupled to the low-pass filter414. An output node of the first differential amplifier 418 is coupledto a non-inverting node of a third differential amplifier 422 and anoutput node of the second differential amplifier 420 is coupled to aninverting node of the third differential amplifier 422. The output nodeof the third differential amplifier 422 is coupled in a negativefeedback loop to the first auxiliary variable current source 408 and tothe second auxiliary variable current source 416.

In an illustrative implementation, the first differential amplifier 418may determine a voltage drop of the first impedance 402 and provide acorresponding output signal to the third differential amplifier 422. Thesecond differential amplifier 420 may determine a voltage drop of thesecond impedance 410 and provide a corresponding output signal to thethird differential amplifier 422. The third differential amplifier 422may determine whether or not there is a difference between the voltagedrop of the first impedance 402 and the voltage drop of the secondimpedance 410. The output signal of the third differential amplifier 422may drive the first auxiliary variable current source 408 and the secondauxiliary current source 416. For example, when there is a differencebetween the voltage drop of the first impedance 402 and the voltage dropof the second impedance 410, the third differential amplifier 422 maydrive the first auxiliary variable current source 408, the secondauxiliary variable current source 416, or a combination thereof, toproduce a compensating current such that the voltage drop of the firstimpedance 402 and the voltage drop of the second impedance 410 areadjusted to be approximately equal.

Further, the monitoring circuit 400 may be implemented with oneauxiliary variable current source, rather than the first auxiliaryvariable current source 408 and the second auxiliary variable currentsource 416 shown in FIG. 4. The single auxiliary variable current sourcemay be coupled to the first impedance 402 or the second impedance 410.The single auxiliary variable current source may provide a compensatingcurrent to the first impedance 402 or the second impedance 410 based onthe output signal of the third differential amplifier 422 when thevoltage drop of the first impedance 402 and the voltage drop of thesecond impedance 410 are different, such that the voltage drop of thefirst impedance 402 and the voltage drop of the second impedance 410 areadjusted to be approximately equal.

FIG. 5 is a schematic diagram of a monitoring circuit 500 of a class ABamplifier output stage circuit arrangement, such as the class ABamplifier output stage circuit arrangement 200 of FIG. 2. The monitoringcircuit 500 may be configured to offset mismatch that occurs in theclass AB amplifier output stage circuit arrangement.

The monitoring circuit 500 includes a first impedance 502 of a class ABamplifier output stage. In some implementations, the first impedance 502may correspond to the first impedance 208 of FIG. 2. A first node of thefirst impedance 502 may be coupled to a low-pass filter 504 and a secondnode of the first impedance 502 may be coupled to a low-pass filter 506.In addition, the first node and the second node of the first impedance502 are coupled to a first auxiliary variable current source 508. Insome implementations, the first auxiliary variable current source 508may be connected in parallel with the first current source 210 of FIG.2, while in other implementations the first auxiliary variable currentsource 508 and the first current source 210 of FIG. 2 may represent thesame current source.

The monitoring circuit 500 also includes a second impedance 510 of aclass AB amplifier output stage. In some implementations, the secondimpedance 510 may correspond to the second impedance 222 of FIG. 2. Afirst node of the second impedance 510 may be coupled to a low-passfilter 512 and a second node of the second impedance 510 may be coupledto a low-pass filter 514. The first node and the second node of thesecond impedance 510 are also coupled to a second auxiliary variablecurrent source 516. In some implementations, the second auxiliaryvariable current source 516 may be connected in parallel with the secondcurrent source 224 of FIG. 2, while in other implementations the secondauxiliary variable current source 516 and the second current source 224of FIG. 2 may represent the same current source. Further, in particularimplementations, the low-pass filters 504, 506, 512, 514 may not benecessary.

Additionally, the monitoring circuit 500 includes a differentialdifference amplifier 518. A first non-inverting node of the differentialdifference amplifier 518 is coupled to the low-pass filter 506 and afirst inverting node of the differential difference amplifier 518 iscoupled to the low-pass filter 504. Further, a second inverting node ofthe differential difference amplifier 518 is coupled to the low-passfilter 512 and a second non-inverting node of the differentialdifference amplifier 518 is coupled to the low-pass filter 514. Anoutput node of the differential difference amplifier 518 is coupled in anegative feedback loop to the first auxiliary variable current source508 and to the second auxiliary variable current source 516.

In an illustrative implementation, the differential difference amplifier518 may determine a voltage drop of the first impedance 502 and avoltage drop of the second impedance 510. The differential differenceamplifier 518 may also determine whether or not there is a differencebetween the voltage drop of the first impedance 502 and the voltage dropof the second impedance 510. The output signal of the differentialdifference amplifier 518 may drive the first auxiliary variable currentsource 508 and the second auxiliary current source 516. For example,when there is a difference between the voltage drop of the firstimpedance 502 and the voltage drop of the second impedance 510, thedifferential difference amplifier 518 may drive the first auxiliaryvariable current source 508, the second auxiliary variable currentsource 516, or a combination thereof, to produce a compensating currentsuch that the voltage drop of the first impedance 502 and the voltagedrop of the second impedance 510 are adjusted to be approximately equal.

FIG. 6 is a flow diagram of a method 600 of correcting mismatch in aclass AB amplifier output stage. The method 600 may utilize a monitoringcircuit to correct mismatch in the class AB amplifier output stage. Theclass AB amplifier output stage may be the class AB amplifier outputstage 200 illustrated in FIG. 2 and the monitoring circuit may be themonitoring circuit 300 shown in FIG. 3, the monitoring circuit 400 ofFIG. 4, or the monitoring circuit 500 of FIG. 5. The class AB amplifieroutput stage and the monitoring circuit may be included in a high-speedamplifier.

Specifics of exemplary methods are described below. However, it shouldbe understood that certain acts need not be performed in the orderdescribed, and may be modified, and/or may be omitted entirely,depending on the circumstances. Moreover, the acts described may beimplemented by a computer, processor or other computing device based oninstructions stored on one or more computer-readable media. Thecomputer-readable media can be any available media that can be accessedby a computing device to implement the instructions stored thereon.

At 602, the method 600 includes generating an output signal of a firstamplifier circuit, such as an operational transconductance amplifier(OTA) or a differential amplifier, based on a voltage drop of a firstimpedance of a class AB amplifier output stage. The first impedance maybe the impedance 208 of FIG. 2 and/or the impedance 302 of FIG. 3, orthe impedance 402 of FIG. 4. In addition, the first amplifier circuitmay be the first OTA 318 of FIG. 3 or the first differential amplifier418 of FIG. 4. At 604, a second amplifier circuit, such as an OTA or adifferential amplifier, generates an output signal based on a voltagedrop of a second impedance of a class AB amplifier output stage. Thesecond impedance may be the impedance 222 of FIG. 2 and/or the impedance310 of FIG. 3, or the impedance 410 of FIG. 4. The second amplifiercircuit may be the second OTA 320 of FIG. 3 or the second differentialamplifier 420 of FIG. 4. In some implementations, the first amplifiercircuit and the second amplifier circuit may represent the functionalityof the differential difference amplifier 518 of FIG. 5 and the impedance502 of FIG. 5 may represent the first impedance and the impedance 510 ofFIG. 5 may represent the second impedance.

At decision 606, the method 600 includes determining whether the voltagedrop of the first impedance and the voltage drop of the second impedanceare different. When there is not a difference between the voltage dropof the first impedance and the voltage drop of the second impedance, themethod returns to 602. When there is a difference between the voltagedrop of the first impedance and a voltage drop of the second impedance,the method advances to 608. At 608, a compensating current is generatedsuch that the voltage drop of the first impedance and the voltage dropof the second impedance are adjusted to be approximately equal. Thecompensating current may be generated by one or more current sources ofthe monitoring circuit. In implementations where the first amplifiercircuit and the second amplifier circuit are OTAs, the one or morecurrent sources may be driven by the output currents of the OTAs.Further, in implementations where the first amplifier circuit and thesecond amplifier circuit are differential amplifiers, the output signalsof the differential amplifiers may be fed into an additionaldifferential amplifier that produces an output signal to drive the oneor more current sources.

CONCLUSION

For the purposes of this disclosure and the claims that follow, theterms “coupled” and “connected” have been used to describe how variouselements interface. Such described interfacing of various elements maybe either direct or indirect. Although the subject matter has beendescribed in language specific to structural features and/ormethodological acts, it is to be understood that the subject matterdefined in the appended claims is not necessarily limited to thespecific features or acts described. Rather, the specific features andacts are disclosed as preferred forms of implementing the claims. Thespecific features and acts described in this disclosure and variationsof these specific features and acts may be implemented separately or maybe combined.

1. An apparatus comprising: a first operational transconductanceamplifier to determine a voltage drop of a first impedance; a secondoperational transconductance amplifier to determine a voltage drop of asecond impedance and wherein an output node of the second operationaltransconductance amplifier is coupled to an output node of the firstoperational transconductance amplifier; and at least one current sourceto adjust the voltage drop of the first impedance and the voltage dropof the second impedance and to provide a compensating current based on adifference between the voltage drop of the first impedance and thevoltage drop of the second impedance.
 2. The apparatus of claim 1,wherein the voltage drop of the first impedance is different from thevoltage drop of the second impedance.
 3. The apparatus of claim 1,wherein the at least one current source is a first variable currentsource, a second variable current source, or a combination thereof, thefirst variable current source, the second variable current source, orthe combination thereof generates the compensating current based on thedifference between the voltage drop of the first impedance and thevoltage drop of the second impedance, the first variable current sourcecoupled to the first operational transconductance amplifier and thesecond variable current source coupled to the second operationaltransconductance amplifier.
 4. The apparatus of claim 1, wherein thefirst impedance and the second impedance are included in a class ABamplifier output stage.
 5. The apparatus of claim 4, wherein the classAB amplifier output stage is included in a high speed amplifier.
 6. Theapparatus of claim 1, wherein a non-inverting input node of the firstoperational transconductance amplifier is coupled to a first low-passfilter and an inverting input node of the first operationaltransconductance amplifier is coupled to a second-low pass filter, andwherein a non-inverting input node of the second operationaltransconductance amplifier is coupled to a third low-pass filter and aninverting input node of the second operational transconductanceamplifier is coupled to a fourth low-pass filter.
 7. The apparatus ofclaims 1, wherein an output node of the first operationaltransconductance amplifier is coupled to a first variable current sourceand an output node of the second operational transconductance amplifieris coupled to a second variable current source.
 8. An apparatuscomprising: a differential amplifier arrangement coupled to a firstimpedance of an amplifier output stage and coupled to a second impedanceof the amplifier output stage, wherein an output node of thedifferential amplifier arrangement is coupled to a variable currentsource arrangement and one or more input nodes of the differentialamplifier arrangement are coupled to the first impedance and the secondimpedance, the differential amplifier arrangement to determine whetherthere is a difference between a voltage drop of the first impedance anda voltage drop of the second impedance.
 9. The apparatus of claim 8,wherein the differential amplifier arrangement comprises a differentialdifference amplifier.
 10. The apparatus of claim 8, wherein thedifferential amplifier arrangement comprises a first differentialamplifier coupled to the first impedance and a second differentialamplifier coupled to the second impedance, and wherein an output node ofthe first differential amplifier is coupled to a non-inverting inputnode of a third differential amplifier and an output node of the seconddifferential amplifier is coupled to an inverting input node of thethird differential amplifier.
 11. The apparatus of claim 10, wherein anoutput node of the third differential amplifier is coupled to thevariable current source arrangement.
 12. The apparatus of claim 8,wherein the variable current source arrangement includes a firstvariable current source coupled to the first impedance and a secondvariable current source coupled to the second impedance.
 13. A methodcomprising: generating an output signal of a first amplifier circuitbased on a voltage drop of a first impedance; generating an outputsignal of a second amplifier circuit based on a voltage drop of a secondimpedance; and generating a compensating current when the voltage dropof the first impedance is different from the voltage drop of the secondimpedance, the compensating current to adjust the voltage drop of thefirst impedance, the voltage drop of the second impedance, or acombination thereof, such that the voltage drop of the first impedanceand the voltage drop of the second impedance are to be adjusted to beapproximately equal.
 14. The method of claim 13, wherein the first andsecond impedances are associated with a class AB amplifier output stage.15. The method of claim 14, wherein the class AB amplifier output stagecomprises a negative channel metal oxide semiconductor (NMOS) transistorcurrent mirror arrangement and a positive channel metal oxidesemiconductor (PMOS) transistor current mirror arrangement.
 16. Themethod of claim 14, wherein a PMOS transistor of the PMOS transistorcurrent mirror arrangement delivers an increased output current byproducing an increased source to drain current that provides a currentpath between a positive supply voltage and an output terminal of theclass AB amplifier output stage when a first input current source, asecond input current source, or a combination thereof, produces anegative input current.
 17. The method of claim 15, wherein an outputcurrent of an NMOS transistor of the NMOS current mirror arrangementdecreases when the first input current source, the second input currentsource, or the combination thereof, produces the negative input current.18. The method of claim 15, wherein an NMOS transistor of the NMOScurrent mirror arrangement delivers an increased output current and anoutput current of a PMOS transistor of the PMOS current mirrorarrangement decreases when a first input current source, a second inputcurrent source, or a combination thereof, produces a positive inputcurrent.
 19. The apparatus of claim 8, wherein the amplifier outputstage is a class AB amplifier output stage.